Bit log likelihood ratio evaluation

ABSTRACT

A system and method are provided for generating bit log likelihood ratio (LLR) values for two-layered Quadrature Phase-Shift Keying (QPSK) turbo decoding in a wireless communications user terminal (UT). The method includes receiving a two-layered QPSK signal with an energy ratio that is unknown, but typically defined as either k1 2  or k2 2 . The method selects a mismatched energy ratio (k 2 ) between k1 2  and k2 2 , and generating bit LLR values for two-layered QPSK turbo decoding, using the mismatched k 2  energy ratio. For example, if the received two-layered QPSK signal is known to have an energy ratio of about 4 or about 6.25. Then, k 2  is selected to be about 5.0625. Alternately stated, the mismatched k 2  energy ratio in selected by determining the approximate midpoint between k1 2  and k2 2 .

CLAIM OF PRIORITY UNDER 35 U.S.C. §119

The present Application for Patent claims priority to ProvisionalApplication No. 60/643,263, entitled LAYERED MODULATION, filed Jan. 11,2005, and assigned to the assignee hereof and hereby expresslyincorporated by reference herein.

FIELD

This invention generally relates to digital communication formats and,more particularly, to a system and method for efficiently generating bitlog likelihood ratio (LLR) values for two-layered Quadrature Phase-ShiftKeying (QPSK) turbo decoding, using a mismatched energy ratio.

BACKGROUND

Wireless communication systems are continually striving to increase thedata bandwidth so that information can quickly be exchanged betweendevices coupled to the communication system. Some of the parameters thatlimit the data bandwidth available to devices include the spectralbandwidth allocated to the devices and the quality of the channellinking the devices.

Wireless communication systems compensate for the various constraints ondata bandwidth using a variety of techniques. A wireless communicationsystem may incorporate multiple encoding techniques, and may select anencoding technique based on a data rate supported by a channel. In sucha system, the communicating devices may negotiate a data rate based onthe capabilities of the channel. Such a communication system may beadvantageous for multiple point-to-point links, but may be less thanideal in a distributed broadcast system where a single transmitterprovides substantially the same data to multiple receivers.

Wireless communication systems may incorporate hierarchical modulation,also referred to as layered modulation, where multiple data streams aresimultaneously transmitted across a hierarchy of data layers. Themultiple data streams can include a base layer that is a robustcommunication link capable of successful reception in nearly allreceiver operating conditions. The multiple data streams can alsoinclude an enhancement layer that is broadcast at a data rate that islower, the same, or at a higher data rate than the base layer. Thecommunications over the enhancement layer may require a higher signalquality at the receiver compared to the base layer. Therefore, theenhancement layer may be more sensitive to variations in the quality ofthe channel.

The receiver is typically ensured the ability to communicate at the baselevel, and can typically demodulate data on the base layer. In channelconditions sufficient to support the enhancement layer, the receiver isalso able to demodulate additional data modulated on the enhancementlayer to provide a higher quality of service or to provide additionaldata bandwidth.

The use of hierarchically modulated signals substantially complicatesthe receiver operation. Further, the receiver may be a portable receiverthat has limited power capacity or limited processing capabilities. Thecomplications to the receiver arising from the incorporation of layeredmodulation operate in contrast to efforts to reduce the size, powerconsumption, and cost of a receiver.

SUMMARY

A decoder for a layered modulation system can be configured toindependently and concurrently decode each of a base and enhancementlayer. The base layer decoder and enhancement layer decoder can beconfigured substantially in parallel and can each operate concurrentlyon the same received layered modulation symbol. Each of the base andenhancement layer decoders can be configured with a bit metric modulethat is configured to determine a signal quality metric based on thereceived symbol. In systems having turbo encoded data, the bit metricmodule can be configured to determine a log likelihood ratio (LLR). Theratio is based in part on a channel estimate and an energy ratio used inthe layered modulation constellation.

Accordingly, a method is provided for generating bit LLR values fortwo-layered QPSK turbo decoding in a wireless communications userterminal (UT). The method includes receiving a two-layered QPSK signalwith an energy ratio that is unknown, and typically defined as eitherk1² or k2². The method selects a mismatched energy ratio (k²) betweenk1² and k2², and generates bit log likelihood ratio (LLR) values fortwo-layered QPSK turbo decoding, using the mismatched k² energy ratio.For example, if the received two-layered QPSK signal is known to have anenergy ratio of either about 4 or about 6.25. Then, k² is selected to beequal to about 5.0625. Alternately stated, the mismatched k² energyratio in selected by determining the approximate midpoint between k1²and k2².

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of an embodiment of a wirelesscommunication system incorporating hierarchical modulation.

FIGS. 2A and 2B are constellation diagrams of a hierarchical modulation.

FIG. 3 is a functional block diagram of an embodiment of a transmitterin a hierarchical coded modulation system.

FIG. 4 is a functional block diagram of an embodiment of a receiverconfigured for operation in a hierarchical modulation system.

FIG. 5 is a detailed depiction of a two-layered QPSK constellationdiagram showing variations in the energy ratio.

FIG. 6 is a flowchart illustrating a method for generating bit LLRvalues for two-layered QPSK turbo decoding in a wireless communicationsuser terminal (UT).

FIG. 7 is a flowchart showing the method of FIG. 6 with additiondetails.

FIG. 8 is a detailed depiction showing an exemplary calculation of v.

FIG. 9 is a detailed depiction showing an exemplary calculation of w.

FIG. 10 is a detailed depiction showing an exemplary calculation ofk·C*·C, or Y.

FIG. 11 is a detailed depiction of an exemplary calculation of b2 andb0.

FIG. 12 is a detailed depiction of an exemplary calculation of b3 andb1.

FIG. 13 is a schematic block diagram of a wireless communications UT,for generating bit LLR values for two-layered QPSK turbo decoding.

FIG. 14 is a schematic block diagram of a processor for generating bitLLR values for two-layered QPSK turbo decoding.

FIG. 15 is a schematic block diagram depicting another variation of awireless communications UT for generating bit LLR values for two-layeredQPSK turbo decoding.

DETAILED DESCRIPTION OF THE INVENTION

A receiver and decoders in a receiver can be configured to decodehierarchical or layered modulation data. The receiver operation andprocessing load is simplified because a base layer decoder can beconfigured to operate substantially in parallel with an enhancementlayer decoder. The base layer and enhancement layer decoders can beconfigured to operate concurrently on the same received symbol in alayered modulation constellation. The enhancement layer decoder canoperate substantially independent of the base layer decoder and does notrely on the results from the base layer decoder when decoding theenhancement layer.

The receiver can be configured to decode hierarchically modulated datathat has been turbo encoded. In such an embodiment, the receiver caninclude a base layer decoder and enhancement layer decoder configuredsubstantially in parallel. Each of the base layer decoder andenhancement layer decoder can include a bit metric module that can beconfigured to determine a signal quality metric, such as a loglikelihood ratio.

The log likelihood ratio values are based, at least in part, on areceived signal and a channel estimate. The bit metric modules can beconfigured to compare channel estimates against a predeterminedthreshold value to determine if the actual channel estimate or apredetermined value is to be used in the determination of the LLRvalues. The receiver operation can be simplified by using the samechannel estimate threshold value for both the base layer and enhancementlayer LLR determination. Different channel estimate thresholds can beused based on different layered modulation energy ratios.

FIG. 1 is a functional block diagram of an embodiment of a wirelesscommunication system 100 incorporating hierarchical modulation,alternatively referred to as layered modulation. The system includes oneor more fixed elements that can be in communication with a user terminal110. The user terminal 110 can be, for example, a wireless telephoneconfigured to operate according to one or more communication standardsusing hierarchical coded modulation (i.e., two-layered QPSK). Forexample, the user terminal 110 can be configured to receive wirelesstelephone signals from a first communication network and can beconfigured to receive data and information from a second communicationnetwork. In some embodiments, both communication networks can implementhierarchical coded modulation, while in other embodiments, one of thecommunication networks may implement hierarchical coded modulation.

The user terminal 110 can be a portable unit, a mobile unit, or, astationary unit. The user terminal 110 may also be referred to as amobile unit, a mobile terminal, a mobile station, user equipment, aportable, a phone, and the like. Although only a single user terminal110 is shown in FIG. 1, it is understood that a typical wirelesscommunication system 100 has the ability to communicate with multipleuser terminals 110.

The user terminal 110 typically communicates with one or more basestations 120 a or 120 b, here depicted as sectored cellular towers. Theuser terminal 110 will typically communicate with a base station, forexample 120 b, which provides the strongest signal strength at areceiver within the user terminal 110.

Each of the base stations 120 a and 120 b can be coupled to a BaseStation Controller (BSC) 140 that routes the communication signals toand from the appropriate base stations 120 a and 120 b. The BSC 140 iscoupled to a Mobile Switching Center (MSC) 150 that can be configured tooperate as an interface between the user terminal 110 and a PublicSwitched Telephone Network (PSTN) 150. The MSC can also be configured tooperate as an interface between the user terminal 110 and a network 160.The network 160 can be, for example, a Local Area Network (LAN) or aWide Area Network (WAN). In one embodiment, the network 160 includes theInternet. Therefore, the MSC 150 is coupled to the PSTN 150 and network160. The MSC 150 can also be coupled to one or more media source 170.The media source 170 can be, for example, a library of media offered bya system provider that can be accessed by the user terminal 110. Forexample, the system provider may provide video or some other form ofmedia that can be accessed on demand by the user terminal 110. The MSC150 can also be configured to coordinate inter-system handoffs withother communication systems (not shown).

In one embodiment, the base stations 120 a and 120 b can be configuredto transmit hierarchically coded signals to the user terminal 110. Forexample, the base stations 120 a and 120 b can be configured to transmita multicast signal that can be directed to the user terminal 110, aswell as other receivers (not shown). The hierarchical coded signals caninclude a base layer signal that is configured to be robust, and anenhancement layer signal that operates at a lower link margin, and as aresult, that is more sensitive to variations in the channel. Theenhancement layer can be configured to provide supplemental data to thedata supplied on the base layer or provide independent data that has alower quality of service requirement.

The wireless communication system 100 can also include a broadcasttransmitter 180 that is configured to transmit a modulatedhierarchically coded signal to the user terminal 110. In one embodiment,the broadcast transmitter 180 can be associated with the base stations120 a and 120 b. In another embodiment, the broadcast transmitter 180can be distinct from, and independent of, the wireless telephone systemcontaining the base stations 120 a and 120 b. The broadcast transmitter180 can be, but is not limited to, an audio transmitter, a videotransmitter, a radio transmitter, a television transmitter, and the likeor some combination of transmitters.

The broadcast transmitter 180 can be configured to receive data from abroadcast media source 182 and can be configured to hierarchically codethe data, modulate a signal based on the hierarchically coded data, andbroadcast the modulated hierarchically coded data to a service areawhere it can be received by the user terminal 110. The broadcasttransmitter 180 can generate, for example, base layer data andenhancement layer data from data received from the broadcast mediasource 182.

The hierarchical coded data configuration can be advantageous becausethe enhancement layer does not carry data that is redundant to thatcarried on the base layer. Additionally, the inability of the receiverto decode the enhancement layer may not result in loss of service. Forexample, the base layer can be configured to deliver video at a standardvideo resolution, and the enhancement layer can provide additional datathat increases the resolution or signal-to-noise ratio (SNR) of thereceived video signal. In another embodiment, the base layer can beconfigured to provide a signal having a predetermined quality, such as avideo signal at 15 frames per second, and the enhancement layer can beconfigured to supplement the information carried on the base layer. Forexample, the enhancement layer can be configured to carry informationused to support a video signal at 30 frames per second. In such aconfiguration, the inability to decode the enhancement layer dataresults in lower resolution signal, lower signal quality, or SNR, butnot a complete loss of signal.

The user terminal 110 can be configured to demodulate the receivedsignal and decode the base layer. The receiver in the user terminal 110can implement error control mechanisms as a standard part of the baselayer decoder. The receiver in the user terminal 110 can use the errorcontrol mechanisms of the base layer decoder to determine a probabilityof successful enhancement layer decoding. The receiver in the userterminal 110 can then determine whether to decode the enhancement layerbased on statistics or metrics generated in the error control mechanismsused in the base layer decoding.

In another embodiment, the user terminal 110 can be configured tosubstantially decode the base layer and enhancement layers concurrently,without relying on base layer information when decoding the enhancementlayer. For example, the user terminal 110 can be configured to determinea single decoder threshold value and use the single decoder thresholdvalue when decoding both the base and enhancement layer. The decoderthreshold can be based in part on a characteristic of the hierarchicallymodulated data. For example, the decoder threshold can be based on aratio of the power or energy of the enhancement layer relative to thebase layer. The decoder threshold can also be based in part on a desirederror rate, such as a symbol error rate, bit error rate, packet errorrate, or frame error rate. The decoder threshold can be fixed or mayvary based, for example, on varying desired quality of service orvarying characteristics of the hierarchically modulated data.

FIGS. 2A and 2B are constellation diagrams of a hierarchical modulation.As an example, the wireless communication system 100 of FIG. 1 mayimplement hierarchical modulation in the manner shown in FIG. 2A. Thehierarchical modulation implementation can be referred to as QuadraturePhase-Shift Keying (QPSK) on QPSK. The implementation includes a QPSKmodulated base layer. Although a QPSK on QPSK hierarchical modulationimplementation is illustrated in FIG. 2A, the decoder apparatus andmethods disclosed herein are not limited to any particular type ofhierarchical modulation. For example, other hierarchical modulationembodiments may use 16-QAM over QPSK, or some other form of hierarchicalmodulation.

The QPSK base layer is defined by four points 202 a-202 d. However, asdescribed later, the points do not need to correspond to actualconstellation points in the hierarchical modulation. The enhancementlayer is also QPSK modulated. The QPSK modulated enhancement layeroccurs on top of the QPSK base layer constellation. The QPSKconstellation for the enhancement layer includes four positions, but theconstellation can be centered about any of the four constellation points202 a-202 d of the base layer.

As an example, a base layer point 202 b occurs in the second quadrant,where the in-phase (I) signal component is negative and the quadrature(Q) signal component is positive. On top of the base layer point 202 bare four constellation points 210 a-210 d of the enhancement layer.Similarly, each quadrant, corresponding to a point 202 a-202 d of thebase layer, has four constellation points of the enhancement layer.

The base and enhancement layer data can be mapped to a constellationsymbol based on a predetermined map or algorithm. For example, the baselayer data and enhancement layer data can each include two bits persymbol, such that the combination of the base layer and enhancementlayer data is four bits. The mapping operation can take the four bitsand map them to a symbol from a predetermined constellation, such as a16-QAM constellation or a QPSK on QPSK constellation.

FIG. 2B is a constellation diagram 260 of an embodiment of a particularhierarchical modulation implementation. The constellation diagram 260 ofFIG. 2B is substantially a 16-QAM constellation in which the base layerdata maps to a particular quadrant of the constellation, and theenhancement layer data maps to the particular position within theconstellation. The 16-QAM constellation 260 does not need to beconsistently spaced, but may be modified to have a consistent spacingwithin each quadrant and a distinct spacing between the nearest pointswithin different quadrants. Furthermore, some of the points in theconstellation may be mirrored with respect to a midpoint in thequadrant.

The input to a signal mapping block includes 2 bits from the base layer(b₃ b₁) and 2 bits from the enhancement layer (b₂ b₀). The base layerstream is transmitted at a higher power level with respect to theenhancement layer stream and the energy ratio k²

The same energy ratio can be used for multiple tones in the same logicalchannel of an OFDM system, where a logical channel can include one ormore tones from the OFDM group of tones. However, the energy ratio canchange from logical channel to logical channel. Therefore, the signalmapping block can map the same data to different constellationsdepending on the energy ratio, with the constellation determined by theenergy ration.

For example, a signal mapping block can be configured to map base andenhancement layer data to one of two constellations, where the twoconstellations correspond to energy ratios of 4 and 9. Note, the layeredmodulation signal constellation follows the Gray mapping, and the signalconstellation for layered modulation is equivalent to the signalconstellation of 16-QAM when the energy ratio, k², is equal to 4.

In other embodiments, the signal constellation for layered modulation isa simple addition of two scaled QPSK signal constellation. Such a simpleadditions of QPSK constellations does not follow a Gray mapping rule asdoes the constellation shown in FIG. 2B. A signal constellation thatdoes not follow Gray mapping may provide reduced performance compared toa constellation conforming to Gray mapping.

The underlying data defining the respective quadrants of the base andenhancement layers can be encoded using one or more encoding processes.The encoding process used can be any encoding process, and the type ofencoding is not a limitation on the decoding apparatus and methodsdisclosed herein, except where the decoder is specific to a particularencoder. The encoder can include, for example, a convolutional encoder,a turbo encoder, a block encoder, an interleaver, a CRC encoder, acombination of encoders, and the like, or some other process orapparatus for encoding data.

FIG. 3 is a functional block diagram of an embodiment of a transmitter300 configured for a hierarchical coded modulation system. In oneembodiment, the transmitter 300 can be implemented in the broadcasttransmitter of the system of FIG. 1. The transmitter 300 of FIG. 3 canbe configured for hierarchical modulation in an Orthogonal FrequencyDivision Multiple Access (OFDMA) or Orthogonal Frequency DivisionMultiplex (OFDM) system using the constellation of FIG. 2B. However, thetransmitter 300 shown in FIG. 3 represents an embodiment and is not alimitation on the disclosed decoder apparatus and methods. For example,a single carrier system can be modulated with hierarchically coded data,and the corresponding decoder in a receiver can be configured to operateon a single carrier with layered modulation.

The transmitter 300 can include substantially similar base layer andenhancement layer processing blocks, 310 and 320, respectively. The baselayer processing block 310 can be configured to process base layer datainto a desired modulation format, for example QPSK. The enhancementlayer processing block 320 can be similarly configured to processenhancement layer data into a desired modulation format, for exampleQPSK.

The base layer processing block 310 and the enhancement layer processingblock 320 receive the respective data from a source encoder (not shown),which can be the broadcast media source of FIG. 1. In one embodiment,the base layer data and the enhancement layer data can include videosignals, audio signals, or some combination of video and audio signals.The video/audio signal in the base layer corresponds to the datarequired to reproduce basic quality of service at the receiver. Thevideo/audio signal in the enhancement layer corresponds to theadditional data required to generate more enhanced quality of service atthe receiver. Hence, users capable of decoding two layers (base layerand enhancement layer) can enjoy fully enhanced quality of video/audiosignal while users capable of decoding the base layer can get a minimumquality of video/audio signal.

Within each of the base layer processing block 310 and the enhancementlayer processing block 320, the data is coupled to a Reed Solomonencoder 301 or 311 for block coding. The output of the Reed Solomonencoders 301 and 311 are coupled to respective turbo encoders 303 and313. The turbo encoders 301 and 311 can be configured to turbo encodethe data according to a predetermined encoding rate. The encoding ratecan be fixed or selectable from a plurality of encoder rates. Forexample, the turbo encoders 301 and 311 can independently be configuredto provide a coding rate of 1/3, 1/2, or 2/3.

The turbo encoder 303 and 313 outputs are coupled to respective bitinterleavers 305 and 315 to improve resistance to burst errors. Theoutput of the bit interleavers 305 and 315 are coupled to respectiveslot assignment modules 307 and 317. The slot assignment modules 307 and317 can be configured to time align the encoded symbols with apredetermined time slot, such as an interleaving time slot in a timedivision multiplexed system. The outputs of the slot alignment modules307 and 317 are coupled to respective scramblers 309 and 319. The outputof the scramblers 309 and 319 represent the encoded base layer andenhancement layer symbols.

The symbols from the two layers are combined at a signal mapping block330. The signal mapping block 330 can be configured to map the base andenhancement layer symbols to a particular point in the constellation forthe layered modulation. For example, the signal mapping block 330 can beconfigured to map a pair of base layer symbols along with a pair ofenhancement layer symbols to a single point in the layered modulationconstellation. The signal mapping block 330 can be configured to mapeach logical channel to a constellation having a predetermined energyratio. However, different logical channels can be mapped toconstellations having different energy ratios.

The output of the signal mapping block 330 is coupled to a timeinterleaver 340 that is configured to interleave the mappedconstellation point to a particular logical channel. As describedearlier, the system may implement a time division multiplexconfiguration where a single logical channel is time multiplexed with aplurality of other logical channels. The aggregate of logical channelscan be time interleaved, or otherwise time multiplexed, using apredetermined time multiplex algorithm, such as a round robinassignment.

The output of the time interleaver 340 is coupled to a subcarrierassignment module 350. The subcarrier assignment module can beconfigured to assign one or more tones, frequencies, or subcarriers froman OFDM tone set to each set of time interleaved logical channels. Thesubset of subcarriers assigned to a set of time interleaved logicalchannels can range from one channel to a plurality of subcarriers up toall available subcarriers. The subcarrier assignment module 350 can mapa serial time interleaved set of logical channels to a subset ofsubcarriers according to a predetermined algorithm. The predeterminedalgorithm can be configured to assign the logical channels in apersistent manner, or can be configured to assign subcarriers accordingto a frequency hopping algorithm.

The output of the subcarrier assignment module 350 is coupled to an OFDMsymbol module 360 that is configured to modulate the subcarriers basedon the assigned layered modulation symbol. The modulated OFDMsubcarriers from the OFDM symbol module 360 are coupled to an IFFTmodule 370 that can be configured to generate an OFDM symbol and appendor prepend a cyclic prefix or a predetermined length.

The OFDM symbols from the IFFT module 370 are coupled to a shaping block380 where the OFDM symbols can be shaped, clipped, windowed, orotherwise processed. The output of the shaping block 380 is coupled to atransmit RF processor 390 for conversion to a desired operatingfrequency band for transmission. For example, the output of the transmitRF processor 390 can include or be coupled to an antenna (not shown) forwireless transmission.

FIG. 4 is a functional block diagram of a receiver 400 configured todecode the hierarchical modulated data generated by the transmitter ofFIG. 3. In one embodiment, the receiver 400 can be implemented in theuser terminal of the system of FIG. 1.

The receiver 400 includes a receive RF processor configured to receivethe transmitted RF OFDM symbols, process them and frequency convert themto baseband OFDM symbols or substantially baseband signals. A signal canbe referred to as substantially a baseband signal if the frequencyoffset from a baseband signal is a fraction of the signal bandwidth, orif signal is at a sufficiently low intermediate frequency to allowdirect processing of the signal without further frequency conversion.The OFDM symbols from the receive RF processor 410 are coupled to an FFTmodule 420 that is configured to transform the OFDM symbols to thehierarchically modulated frequency domain subcarriers.

The FFT module 420 can be configured to couple one or more subcarriers,such as predetermined pilot subcarriers, to a channel estimator 430. Thepilot subcarriers can be, for example, one or more equally spaced setsof OFDM subcarriers. The channel estimator 430 is configured to use thepilot subcarriers to estimate the various channels that have an effecton the received OFDM symbols. In one embodiment, the channel estimator430 can be configured to determine a channel estimate corresponding toeach of the subcarriers. The channel estimates at a particularsubcarrier can be used as a channel estimate for adjacent subcarriers,for example, those subcarriers within a predetermined coherencebandwidth of the pilot subcarrier.

The subcarriers from the FFT module 420 and the channel estimates arecoupled to a subcarrier symbol deinterleaver 440. The symboldeinterleaver 440 can be configured to reverse the symbol mappingperformed by the subcarrier assignment module of FIG. 3.

The receiver 400 is configured to perform base layer decoding andenhancement layer decoding on each OFDM subcarrier or tone. FIG. 4illustrates a single base layer decoder and enhancement layer decoderfor the sake of clarity and brevity.

The base layer decoder and enhancement layer decoder can operatesubstantially in parallel. Each of the decoder modules can be configuredto operate concurrently on the same received symbols. The enhancementlayer decoder can thus operate substantially independently of the baselayer decoder and does not rely on the results of the base layer decoderwhen decoding the enhancement layer data.

The decoders illustrated in the receiver 400 embodiment of FIG. 4 areconfigured to decode turbo encoded layered modulation data. Of course,if the transmitter is configured to generate some other type ofencoding, the decoders in the receiver 400 would be matched to theencoder type. For example, the transmitter can be configured to encodethe data using turbo coding, convolutional coding, Low Density ParityCheck (LDPC) coding, or some other encoding type. In such an embodiment,the receiver 400 is configured with the complementary decoders. Thus,each of the base layer decoders and enhancement layer decoders in thereceiver 400 can be configured to provide turbo decoding, convolutionaldecoding, such as using Viterbi decoding, LDPC decoding, or some otherdecoder or combination of decoders.

Each of the hierarchically modulated tones is coupled to a base layerbit metric module 450 and an enhancement layer bit metric module 460.The bit metric modules 450 and 460 can operate on the hierarchicallymodulated tone to determine a metric indicative of the quality of thereceived symbol.

In one embodiment, where the symbols are turbo coded, the bit metricmodules 450 and 460 can be configured to determine a log likelihoodratio (LLR) of the received symbol. The LLR is the logarithm of thelikelihood ratio. The ratio can be defined as the probability that theoriginal bit is 1 over the probability that the original bit is equal to0. Alternatively, the ratio can be defined in a reverse way, where theLLR is the probability that the original bit is 0 over the probabilitythat the original bit is equal to 1. There is no substantial differencebetween these two definitions. The bit metric modules 450 and 460 canuse, for example, the symbol magnitudes and the channel estimate todetermine the LLR values.

Each bit metric module 450 and 460 utilizes a channel estimate and areceived signal to determine a LLR value. A noise estimate may also beused. However, the noise estimate term can be substantially ignored if aturbo decoding method that provides the same results regardless of thenoise estimate is used. In such an embodiment, the bit metric modules450 and 460 hardware can use a predetermined value as the noise estimatein calculating LLR values.

The output of the base bit metric module 450 is coupled to a base layerprocessor 470. The output of the enhancement layer bit metric module 460is coupled to an enhancement layer processor 480 that functionally,operates similarly to the base layer processor 470. For example, the LLRvalues are coupled from the bit metric modules 450 and 460 to therespective base layer or enhancement layer processors 470 and 480.

The base layer processor 470 includes a descrambler 472 configured tooperate on the received LLR values to reverse the symbol scramblingperformed in the encoder. The output of the symbol descrambler 472 iscoupled to a bit interleaver 474 that is configured to deinterleave thepreviously interleaved symbols. The output of the bit deinterleaver 474is coupled to a turbo decoder 476 that is configured to decode turboencoded symbols according to the coding rate used by the turbo encoder.For example, the turbo decoder 476 can be configured to perform decodingof rate 1/3, 1/2, or 2/3 turbo encoded data. The turbo encoder 476operates, for example, on the LLR values. The decoded outputs from theturbo decoder 476 is coupled to a Reed Solomon decoder 478 that can beconfigured to recover the base layer bits based in part on the ReedSolomon encoded bits. The resulting base layer bits are transferred to asource decoder (not shown).

The enhancement layer processor 480 operates similar to the base layerprocessor 470. A descrambler 482 receives the LLR values from theenhancement bit metric module 460. The output is coupled to a bitdeinterleaver 484 and the turbo decoder 486. The output of the turbodecoder 486 is coupled to the Reed Solomon decoder 488. The resultingenhancement layer bits are transferred to a source decoder (not shown).

The exact expression for the LLR is given by:${LLR}_{n} = {{\ln\left( {\sum\limits_{{{G{(S)}}:b_{n}} = 0}{\exp\left( {- \frac{{{r - {{cG}(S)}}}^{2}}{N_{0}}} \right)}} \right)} - {{\ln\left( {\sum\limits_{{{G{(S)}}:b_{n}} = 1}{\exp\left( {- \frac{{{r - {{cG}(S)}}}^{2}}{N_{0}}} \right)}} \right)}.}}$

In the equation, LLR_(n) is the LLR of the n'th bit encoded by themodulation symbol and b_(n) denotes the n'th bit of the constellationpoint G(S). The value r represents the received symbol, c represents thechannel estimate, and N₀ represents the noise estimate. Computing theexact solution is generally too complicated or processing intensive tobe implemented in practice. An approximation can be determined as themaximum of the variables. For QPSK this approximation in factcorresponds to the exact LLR expression. For two-layered QPSK withenergy ratio k², the approximation is described in detail below.

In the LLR calculation block the LLR value depends on a channel estimatefrom the channel estimation block. The performance of each layer dependson a threshold value being used in the channel estimation algorithm. Thechannel estimation threshold value represents a value over which thechannel estimate is used. That is, if the channel estimate exceeds thethreshold value, the actual channel estimate is used. Conversely, if thechannel estimate is less than the threshold value, the channel estimateis assigned a predetermined value, which can be, for example, zero orsome other sufficiently small value. If the channel estimate is equal tothe threshold value, the receiver can be configured to use the actualchannel estimate or use the predetermined value. Either option ispractical, provided the decision is executed consistently.

The channel estimation module in the receiver estimates the channel foreach tone in a multiple channel system, such as an OFDM system. Thus,the channel estimation module or each bit metric module can compare thechannel estimate to the threshold. It may be advantageous to perform thecomparison of the channel estimate to the threshold at the bit metricmodules, because the optimal threshold value can depend on the energyratio.

The threshold can be optimized for the following two channel models;Repeated International Telecommunications Union (ITU) Pedestrian B(PEDB) model with 120 km/hr and repeated Advanced Television SystemsCommittee (ATSC) model with 20 km/hr.

FIG. 5 is a detailed depiction of a two-layered QPSK constellationdiagram showing variations in the energy ratio. Shown are energy ratios(k²) of 4, 5.0625, and 6.25.

Referring again to FIG. 2B, a two-layered QPSK constellation is depictedshowing the relationship between layers with an energy ratio (k²) of 4.The value k can be depicted in the diagram as the Real component a baselayer point, divided by the Real component of the difference between thebase layer point and one of the enhancement layer points.

In some circumstances, a receiver may not be aware of the energy ratio(i.e., 4.0 or 6.25) used by the transmitter. The present inventiondescribes a process for efficiently decoding two-layered QPSK data inthis circumstance. The receiver defines the relationship betweenconstellation layers as in FIG. 2B, even if the energy ratio is unknown,as well as a turbo code rate of 1/3, 1/2, or 2/3. The bit-width of inputand output are 9-bit signed integer (9 s) and 6-bit signed (6 s) integeras required. The decoding metric minimizes the complexity of hardwareimplementation at the receiver side, with only a small performancedegradation, by using a mismatch energy ratio that works for both energyratio extremes. That is, the mismatch energy ratio can be used togenerate bit LLR for the turbo decoder, with a worst case performancedegradation of 0.1 dB for a repeated PEBD channel (120 km/hr) and arepeated ATSC channel (20 km/hr).

FIG. 6 is a flowchart illustrating a method for generating bit LLRvalues for two-layered QPSK turbo decoding in a wireless communicationsuser terminal (UT). Although the method is depicted as a sequence ofnumbered steps for clarity, the numbering does not necessarily dictatethe order of the steps. It should be understood that some of these stepsmay be skipped, performed in parallel, or performed without therequirement of maintaining a strict order of sequence. The method startsat Step 600.

Step 602 receives a two-layered QPSK signal with an energy ratio that istypically about k1² or about k2². Step 604 selects a mismatched energyratio (k²) between k1² and k2². Step 606 generates bit LLR values fortwo-layered QPSK turbo decoding, using the mismatched k² energy ratio.

FIG. 7 is a flowchart showing the method of FIG. 6 with additiondetails. The method starts at Step 700. Step 702 receives a two-layeredQPSK signal with an energy ratio that is unknown, but typically definedas either k1² or k2². Step 704 selects a mismatched energy ratio (k²)between k1² and k2². Step 706 generates bit LLR values for two-layeredQPSK turbo decoding, using the mismatched k² energy ratio. For example,Step 702 may receive a two-layered QPSK signal with an energy ratio ofeither about 4 or about 6.25. Then, Step 704 selects k² to be equal toabout 5.0625. Step 708 turbo decodes using the bit LLR values generatedin Step 706.

In one aspect, selecting the mismatched k² energy ratio in Step 704includes substeps. Step 704 a determines the approximate midpointbetween k1² and k2², and Step 704 b sets k² equal to the approximatemidpoint. Note, the approximate midpoint may be determined on-the-fly,or it may be a predetermined value that is loaded and set in thefactory, or upon initialization of the UT device. The term “approximate”is used to account for errors in the transmission and receivingprocesses, as well as the limitations imposed by using a limited numberof bit places for calculations. In another aspect, Step 706 generatesbit LLR values for two-layered QPSK turbo decoding with a worst-casedegradation of less than 0.1 dB (see Table 1, below).

As described in detail above, Step 702 receives a two-layered QPSKsignal with in phase (I) and quadrature (Q) components per symbol, andStep 706 generates the LLR values for four bits (b3, b2, b1, and b0)using the mismatched k² energy ratio.

More explicitly, since the LLR values for b0 and b2 only depend on theReal component of the complex received signal (r), the LLR values can bedetermined for each b0 and b2 bit as follows:${{LLR}_{2,0} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times \left( {{J} - {k \cdot C^{*} \cdot C}} \right)}};$

-   -   where J is v for b0, and w for b2;    -   v=√{square root over (2(1+k²))}Re[C*r];    -   w=√{square root over (2(1+k²))}Im[C*r];    -   r is the complex received signal;    -   C is the complex channel estimate; and,    -   C* is the complex conjugate of C.

Likewise, the LLR values are determined for each b1 and b3 bit asfollows:${LLR}_{1} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times {{sgn}(J)} \times \left\{ \begin{matrix}{\left( {k - 1} \right){J}} & {if} & {{J} \leq {k \cdot C^{*} \cdot C}} \\{k\left( {{J} - {C^{*} \cdot C}} \right)} & {if} & {{J} > {k \cdot C^{*} \cdot C}}\end{matrix} \right.}$

-   -   where J is v for b1, and w for b3;    -   sgn(J) is the sign bit of J.

FIG. 8 is a detailed depiction showing an exemplary calculation of v.This calculation can be performed in software, hardware, or acombination of software and hardware. As shown, v is determined inresponse to the following operations:

-   -   representing C_(I), C_(Q), r_(I), and r_(Q) as 9-bit signed        integers, and √{square root over (1+k²)} as a 6-bit unsigned        integer;    -   √{square root over (2)}([(C_(I))·r_(I)]+[(C_(Q))·r_(Q)])=M,        saturated to a 17-bit signed integer; and,    -   M·√{square root over (1+k²)}=v, where the result is rounded to        create a 15-bit signed integer.

FIG. 9 is a detailed depiction showing an exemplary calculation of w. Asshown, w can be determined in response to the following operations:

-   -   representing C_(I), C_(Q), r_(I), and r_(Q) as 9-bit signed        integers, and as a 6-bit unsigned integer;    -   √{square root over (2)}([(C_(Q))·r_(Q)]−[(C_(I))·r_(I)])=N,        saturated to a 17-bit signed integer; and,    -   N·√{square root over (1·k²)}=w, where the result is rounded to        create a 15-bit signed integer.

FIG. 10 is a detailed depiction showing an exemplary calculation ofk·C*·C, or Y. As shown, k·C*·C is determined in response to thefollowing operations:

-   -   representing C_(I) and C_(Q) as 9-bit signed integers (a scaling        factor or 2⁷), and k as a 3-bit unsigned integer;    -   [(C_(I))·(C_(I))]+[(C_(Q))·(C_(Q))]=P, a 17-bit unsigned integer        (scaling factor of 2¹⁴); and,    -   P·k=k·C*·C, where the result is rounded to create a 15-bit        unsigned integer Y (scaling factor of 2¹¹).

FIG. 11 is a detailed depiction of an exemplary calculation of b2 andb0. As shown, b2 and b0 (b_(2,0)) are determined in response to thefollowing operations:

-   -   truncating the least significant bit (LSB) of Y, creating a        14-bit unsigned integer YY;    -   taking the absolute value of J (where J is v for b0, and w for        b2), creating a 14-bit unsigned integer U;    -   U−YY=R, a 15-bit signed integer;    -   left-shifting R by 7 bits (multiplying R by 2⁷);    -   rounding the result, creating a 12-bit signed integer DD; and,    -   saturating DD, creating a 6-bit signed integer b_(2,0).

FIG. 12 is a detailed depiction of an exemplary calculation of b3 andb1. As shown, b3 and b1 (b_(3,1)) can be determined in response to thefollowing operations:

-   -   taking the absolute value of J (where J is v for b1, and w for        b3), creating a 14-bit unsigned integer U;    -   U·(k−1)=X, a 16-bit unsigned integer;    -   U·k=X′, a 17-bit unsigned integer;    -   X′−Y, creating an 18-bit signed integer S;    -   (2·U)−Y, creating a 16-bit signed integer T;    -   multiplexing X with S, using the sign bit of T as a select        signal, creating an 18-bit unsigned integer Z;    -   left-shifting Z by 5 bits (multiplying by 2⁵);    -   rounding the result, creating a 12-bit signed integer AA; and,    -   saturating AA, creating a 6-bit signed integer BB;    -   multiplying BB by the most significant bit (the sign) of J,        creating a 6-bit signed integer b_(3,1).

Simulation results from a fixed point implementation show that the worstcase performance degradation due to energy ratio mismatch is 0.1 dB fora repeated PEBD channel (120 km/hr) and a repeated ATSC channel (20km/hr). TABLE 1 Degradation due to mismatch with Rx energy ratio at5.0625 Tx Energy Layer Ratio ATSC ⅓ ATSC ½ ATSC ⅔ PEDB ⅓ PEDB ½ PEDB ⅔Enhan. 4.0   ˜0 dB   0.03 dB   ˜0 dB   ˜0 dB −0.06 dB Floor Enhan. 6.25  ˜0 dB   ˜0 dB Floor   0.05 dB    0.1 dB Floor Base 4.0 −0.05 dB −0.05dB −0.25 dB   ˜0 dB   ˜0 dB 0.05 dB Base 6.25   0.03 dB  −0.1 dB   ˜0 dB  ˜0 dB   ˜0 dB 0.08 dB

In the layered modulation studied, the base layer and enhancement layerwere unequally error protected. The base layer is typically betterprotected than enhancement layer. A negative value in Table 1 means thatthe performance using the mismatched k² is even better than can beobtained using the matched energy ratio.

FIG. 13 is a schematic block diagram of a wireless communications UT,for generating bit LLR values for two-layered QPSK turbo decoding. TheUT 1300 comprises a receiver front end 1302 having an air interfaceinput on line 1304 to accept a two-layered QPSK signal with an energyratio of either k1² or k2², and outputs on lines 1306 and 1308 to supplya complex received signal components (r) and complex channel estimates(C), respectively. A log likelihood ratio (LLR) module 1310 has inputson lines 1306 and 1308 to receive the complex received signal componentsand complex channel estimates. FIG. 13 is a simplified version of thesystem shown in FIG. 4. Elements 410, 420, 430, and 440 of FIG. 4 may beunderstood to perform many of the functions performed by the receiverfront end 1302 of FIG. 13. In some aspects, UT 1300 can be enabled toperform as UT 110 of FIG. 1.

The LLR module 1310 selects a mismatched energy ratio (k²) between k1²and k2², and supplies bit LLR values for two-layered QPSK turbo decodingat an output on line 1312, using the k² energy ratio. Output 1312 isconnected to turbo decoder 1314. Typically, the LLR module 1310 selectsk² as the approximate midpoint between k1² and k2². For example, if thereceiver 1302 accepts a two-layered QPSK signal with an energy ratio ofeither about 4 or about 6.25, then the LLR module 1410 selects k² to beequal to about 5.0625.

As described above in the explanation of FIGS. 7-11, the receiver 1402accepts a two-layered QPSK signal with in phase (I) and quadrature (Q)components, and the LLR module 1410 generates four LLR bit values (b3,b2, b1, and b0).

The LLR module 1310 determines the LLR values for each b0 and b2 bit asfollows:${{LLR}_{2,0} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times \left( {{J} - {k \cdot C^{*} \cdot C}} \right)}};$

-   -   where J is v for b0, and w for b2;    -   v=√{square root over (2(1+k²))}Re[C*r];    -   w=√{square root over (2(1+k²))}Im[C*r];    -   r is the complex received signal;    -   C is the complex channel estimate; and,    -   C* is the complex conjugate of C.

The LLR module 1310 determines the LLR values for each b1 and b3 bit asfollows:${LLR}_{1} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times {{sgn}(J)} \times \left\{ \begin{matrix}{\left( {k - 1} \right){J}} & {if} & {{J} \leq {k \cdot C^{*} \cdot C}} \\{k\left( {{J} - {C^{*} \cdot C}} \right)} & {if} & {{J} > {k \cdot C^{*} \cdot C}}\end{matrix} \right.}$

-   -   where J is v for b1, and w for b3; and,    -   sgn(J) is the sign bit of J.

Additional details of the LLR functions can be found in the explanationof FIGS. 7-11 above, and are not repeated here in the interest ofbrevity.

FIG. 14 is a schematic block diagram of a processor for generating bitLLR values for two-layered QPSK turbo decoding. The processor 1400comprises inputs on line 1402 to receive complex received signals andcomplex channel estimates on line 1404 for QPSK symbols.

A log likelihood ratio (LLR) section 1406 has inputs on lines 1402 and1404 to receive the complex received signals (r) and complex channelestimates (C). The LLR section 1406 selects a mismatched energy ratio(k²) between k1² and k2² and generates bit LLR values for two-layeredQPSK turbo decoding, using the k² energy ratio. Outputs on line 1408 areconnected to a turbo decoder (not shown). Typically, the LLR section1406 selects k² as the approximate midpoint between k1² and k2². Forexample, if processor inputs 1402/1404 accept complex received signaland complex channel estimates for a two-layered QPSK signal with anenergy ratio of either about 4 or about 6.25, the LLR section 1406selects k² to be equal to about 5.0625.

Also shown are a microprocessor (μP) 1410 and memory 1412. Some, or allof the above-mentioned processes may be performed with the aid ofmicroprocessor executable instructions stored in memory 1412. In thatcase, the invention can alternately be described as a signal bearingmedium tangibly embodying a program of machine-readable instructions,stored in memory 1412, executable by a digital processing apparatus,such as microprocessor 1410, to perform operations for generating bitLLR values for two-layered QPSK turbo decoding. The operations includethe steps of: receiving a two-layered QPSK signal with an energy ratioof either k1² or k2²; selecting a mismatched energy ratio (k²) betweenk1² and k2²; and, generating bit LLR values for two-layered QPSK turbodecoding, using the mismatched k² energy ratio.

Selecting the mismatched k² energy ratio may include determining theapproximate midpoint between k1² and k2², and setting k² equal to theapproximate midpoint. For example, if the energy ratio is either about 4or about 6.25, then k² is selected to be equal to about 5.0625.Additional details of the LLR functions can be found in the explanationof FIGS. 7-11 above, and are not repeated here in the interest ofbrevity.

FIG. 15 is a schematic block diagram depicting another variation of awireless communications UT for generating bit LLR values for two-layeredQPSK turbo decoding. The UT 1500 comprises a means 1502 for accepting atwo-layered QPSK signal with an energy ratio of either k1² or k2². TheUT comprises a means 1504 for supplying a complex received signalcomponents (r) and complex channel estimates (C). The UT furthercomprises a means 1506 for selecting a mismatched energy ratio (k²)between k1² and k2², and a means 1508 for supplying bit LLR values fortwo-layered QPSK turbo decoding at an output 1510, using the k² energyratio. Typically, means 1506 selects k² as the approximate midpointbetween k1² and k2². For example, if means 1502 accepts a two-layeredQPSK signal with an energy ratio that is known to be either about 4 orabout 6.25, then means 1506 selects k² to be equal to about 5.0625.

The various illustrative logical blocks, modules, and circuits describedin connection with the UT and processor disclosed herein may beimplemented or performed with a general purpose processor, a digitalsignal processor (DSP), a Reduced Instruction Set Computer (RISC)processor, an application specific integrated circuit (ASIC), a fieldprogrammable gate array (FPGA) or other programmable logic device,discrete gate or transistor logic, discrete hardware components, or anycombination thereof designed to perform the functions described herein.A general purpose processor may be a microprocessor, but in thealternative, the processor may be any processor, controller,microcontroller, or state machine. A processor may also be implementedas a combination of computing devices, for example, a combination of aDSP and a microprocessor, a plurality of microprocessors, one or moremicroprocessors in conjunction with a DSP core, or any other suchconfiguration.

The steps of a method, process, or algorithm described in connectionwith the embodiments disclosed herein may be embodied directly inhardware, in a software module executed by a processor, or in acombination of the two. The various steps or acts in a method or processmay be performed in the order shown, or may be performed in anotherorder. Additionally, one or more process or method steps may be omittedor one or more process or method steps may be added to the methods andprocesses. An additional step, block, or action may be added in thebeginning, end, or intervening existing elements of the methods andprocesses.

A system and method have been provided for decoding a two-layered QPSKsignal with an unknown energy ratio. Examples of specific energy ratiosand exemplary calculation schemes have been presented to illustrate theinvention. However, the invention is not limited to just these examples.Likewise, although a two-layered QPSK signal has been described, thepresent invention system and method are equally applicable tomulti-layered QPSK.

The above description of the disclosed embodiments is provided to enableany person of ordinary skill in the art to make or use the disclosure.Various modifications to these embodiments will be readily apparent tothose of ordinary skill in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the disclosure. Thus, the disclosure is not intendedto be limited to the embodiments shown herein but is to be accorded thewidest scope consistent with the principles and novel features disclosedherein. Other variations and embodiments of the invention will occur tothose skilled in the art.

1. In a wireless communications user terminal (UT), a method forgenerating bit log likelihood ratio (LLR) values for two-layeredquadrature phase-shift keying (QPSK) turbo decoding, the methodcomprising: receiving a two-layered QPSK signal with an energy ratioselected from the group comprising k1² and k2²; selecting a mismatchedenergy ratio (k²) between k1² and k2²; and, generating bit LLR valuesfor two-layered QPSK turbo decoding, using the mismatched k² energyratio.
 2. The method of claim 1 wherein selecting the mismatched k²energy ratio includes: determining the approximate midpoint between k1²and k2²; and, setting k² equal to the approximate midpoint.
 3. Themethod of claim 1 wherein receiving the two-layered QPSK includesreceiving a two-layered QPSK signal with an energy ratio selected fromthe group comprising about 4 and about 6.25; and, wherein selecting themismatched k² energy ratio includes selecting k² to be about 5.0625. 4.The method of claim 3 wherein generating bit LLR values for two-layeredQPSK turbo decoding includes decoding with a worst case degradation ofless than about 0.1 dB.
 5. The method of claim 1 wherein receiving thetwo-layered QPSK includes receiving a two-layered QPSK signal with inphase (I) and quadrature (Q) components per symbol; and whereingenerating bit LLR values for two-layered QPSK turbo decoding includesdetermining LLR values for four bits (b3, b2, b1, and b0).
 6. The methodof claim 5 wherein determining the bit LLR values includes determiningthe LLR values for each b0 and b2 bit as follows:${{LLR}_{2,0} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times \left( {{J} - {k \cdot C^{*} \cdot C}} \right)}};$where J is v for b0, and w for b2; v=√{square root over(2(1+k²))}Re[C*r]; w=√{square root over (2(1+k²))}Im[C*r]; r is thecomplex received signal; C is the complex channel estimate; and, C* isthe complex conjugate of C.
 7. The method of claim 6 wherein determiningthe bit LLR values includes determining the LLR values for each b1 andb3 bit as follows:${LLR}_{1} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times {{sgn}(J)} \times \left\{ \begin{matrix}{\left( {k - 1} \right){J}} & {if} & {{J} \leq {k \cdot C^{*} \cdot C}} \\{k\left( {{J} - {C^{*} \cdot C}} \right)} & {if} & {{J} > {k \cdot C^{*} \cdot C}}\end{matrix} \right.}$ where J is v for b1, and w for b3; sgn(J) is thesign bit of J.
 8. A wireless communications user terminal (UT), forgenerating bit log likelihood ratio (LLR) values for two-layeredquadrature phase-shift keying (QPSK) turbo decoding, the UT comprising:a receiver having an air interface input to accept a two-layered QPSKsignal with an energy ratio selected from the group comprising k1² andk2²; and outputs to supply a complex received signal components andcomplex channel estimates; and, a LLR module having inputs to receivethe complex received signal components and complex channel estimates,the LLR module selecting a mismatched energy ratio (k²) between k1² andk2², and supplying bit LLR values for two-layered QPSK turbo decoding atan output, using the k² energy ratio.
 9. The UT of claim 8 wherein theLLR module selects k² as the approximate midpoint between k1² and k2².10. The UT of claim 8 wherein the receiver accepts a two-layered QPSKsignal with an energy ratio selected from the group comprising about 4and about 6.25; and, wherein the LLR module selects k² to be about5.0625.
 11. The UT of claim 8 wherein the receiver accepts a two-layeredQPSK signal with in phase (I) and quadrature (Q) components; and whereinthe LLR module determines four bit LLR values (b3, b2, b1, and b0). 12.The UT of claim 11 wherein the LLR module determines the LLR values foreach b0 and b2 bit as follows:${{LLR}_{2,0} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times \left( {{J} - {k \cdot C^{*} \cdot C}} \right)}};$where J is v for b0, and w for b2; v=√{square root over(2(1+k²))}Re[C*r]; w=√{square root over (2(1+k²))}Im[C*r]; r is thecomplex received signal; C is the complex channel estimate; and, C* isthe complex conjugate of C.
 13. The UT of claim 12 wherein the LLRmodule determines the LLR values for each b1 and b3 bit as follows:${LLR}_{1} = {\frac{1}{N_{0}} \times \frac{2}{1 + k^{2}} \times {{sgn}(J)} \times \left\{ \begin{matrix}{\left( {k - 1} \right){J}} & {if} & {{J} \leq {k \cdot C^{*} \cdot C}} \\{k\left( {{J} - {C^{*} \cdot C}} \right)} & {if} & {{J} > {k \cdot C^{*} \cdot C}}\end{matrix} \right.}$ where J is v for b1, and w for b3; and, sgn(J) isthe sign bit of J.
 14. A processor for generating bit log likelihoodratio (LLR) values for two-layered quadrature phase-shift keying (QPSK)turbo decoding, the processor comprising: inputs to receive complexreceived signals and complex channel estimates for QPSK symbols; a loglikelihood ratio (LLR) section having inputs to receive the complexreceived signals and complex channel estimates, the LLR sectionselecting a mismatched energy ratio (k²) between k1² and k2² andgenerating bit LLR values for two-layered QPSK turbo decoding using thek² energy ratio; and, outputs connected to a turbo decoder, to supplybit LLR values.
 15. The processor of claim 14 wherein the LLR sectionselects k² as the approximate midpoint between k1² and k2².
 16. Theprocessor of claim 14 wherein the processor inputs accept complexreceived signal and complex channel estimates for two-layered QPSKsignals with an energy ratio selected from the group comprising about 4and about 6.25; and, wherein the LLR section selects k² to be about5.0625.
 17. A signal bearing medium tangibly embodying a program ofmachine-readable instructions executable by a digital processingapparatus to perform operations for generating bit log likelihood ratio(LLR) values for two-layered QPSK turbo decoding, the operationscomprising: receiving a two-layered QPSK signal with an energy ratioselected from the group comprising k1² and k2²; selecting a mismatchedenergy ratio (k²) between k1² and k2²; and, generating bit LLR valuesfor two-layered QPSK turbo decoding, using the mismatched k² energyratio.
 18. The operations of claim 17 wherein selecting the mismatchedk² energy ratio includes: determining the approximate midpoint betweenk1² and k2²; and, setting k² equal to the approximate midpoint.
 19. Theoperations of claim 17 wherein receiving the two-layered QPSK includesreceiving a two-layered QPSK signal with an energy ratio selected fromthe group comprising about 4 and about 6.25; and, wherein selecting themismatched k² energy ratio includes selecting k² to be about 5.0625. 20.A wireless communications user terminal (UT), for generating bit loglikelihood ratio (LLR) values for two-layered quadrature phase-shiftkeying (QPSK) turbo decoding, the UT comprising: a means for accepting atwo-layered QPSK signal with an energy ratio selected from the groupcomprising k1² and k2²; a means for supplying a complex received signalcomponents and complex channel estimates; a means for selecting amismatched energy ratio (k²) between k1² and k2², and, a means forsupplying bit LLR values for two-layered QPSK turbo decoding at anoutput, using the k² energy ratio.
 21. The UT of claim 20 wherein themeans for selecting the mismatched energy ratio selects k² as theapproximate midpoint between k1² and k2².
 22. The UT of claim 20 whereinthe means for accepting the two-layered QPSK signal accepts atwo-layered QPSK signal with an energy ratio selected from the groupcomprising about 4 and about 6.25; and, wherein the means for selectingthe mismatched energy ratio selects k² to be about 5.0625.
 23. Acomputer readable medium having computer executable instructions storedthereon to execute components of a wireless receiver, comprising:receiving a two-layered QPSK signal with an energy ratio selected fromthe group comprising k12 and k22; selecting a mismatched energy ratio(k2) between k12 and k22; and, generating bit LLR values for two-layeredQPSK turbo decoding, using the mismatched k2 energy ratio.